The invention relates to digital communication employing Orthogonal Frequency Division Multiplexing (OFDM), and more particularly to using properties of the guard interval to determine initial timing synchronization.
Orthogonal Frequency Division Multiplexing (OFDM) is a method that has been increasingly popular for transmitting digital information. Currently it is, for example, used for Digital Audio Broadcasting (DAB), Digital Video Broadcasting (DVB), and for some Wireless Local Area Network (WLAN) standards like IEEE 802.11a and IEEE 802.1g. One of the reasons for using OFDM is that it allows for communication over highly time-dispersive channels using reasonable complexity at the receiver side.
The way to handle large delay spreads for a system based on OFDM is to make use of a guard interval (GI). The GI (also referred to in the literature as a “cyclic prefix”, or “CP”) is simply a copy of the last part of an OFDM symbol that is sent before the actual symbol. This is schematically illustrated in FIG. 1, which shows a number of symbols. An exemplary one of the symbols 101 includes a last portion 103 that is transmitted as a preceding guard interval 105 (time flows from left to right in the figure). Other guard intervals are similarly formed from end portions of their immediately succeeding symbols.
It is well-known that for a system based on OFDM the effect of the time-dispersive channel, known as inter-symbol interference (ISI), can be avoided provided that the length of the GI, TG, is at least as long as the (maximum) duration of the impulse response of the channel, henceforth denoted Tm. Because of the ability of an OFDM system to handle large delay spreads, it is very suitable for so-called Single Frequency Networks (SFN), which might be used for broadcasting. (In a single frequency network, geographically spaced transmitters operate on a same frequency. To reduce interference, they are time synchronized with one another.)
Now, as discussed above, ISI free reception is possible whenever Tm≦TG. However, this requires identifying the start of the information carrying part of the signal. For this reason, OFDM receivers include arrangements for estimating the timing and frequency of the received signal. FIG. 2 is a block diagram of an exemplary OFDM receiver. An analog signal, r(t), generated by receiving and downconverting a radiofrequency signal, is supplied to an analog-to-digital (A/D) converter 201. The digitized signal, r(k), is then supplied to a coarse timing and frequency estimation unit 203, which generates a coarse estimate of the timing and frequency offset of the received signal. (The frequency offset is the difference between the frequency of the transmitted signal and the frequency of the received signal.) This information is supplied to a frequency correction unit 205 as well as a GI removal unit 207. The GI removal unit 207 also receives the output of the frequency correction unit 205. Based on the best timing and frequency information available, the GI removal unit 207 removes the GI and supplies the information part of the received signal to an FFT unit 209, whose output is supplied to the remainder of the receiver, including a refined timing and frequency estimation unit 211, which is able to generate more accurate timing and frequency information from the FFT output signal. The more accurate frequency information is fed back to the frequency correction unit 205 to improve the receiver's performance. The more accurate timing information is similarly fed back to the GI removal unit 207 to improve the receiver's performance.
Focusing now on the coarse timing and estimation unit 203, the usual way to find the start of the symbol is by correlating the received signal with a delayed and complex conjugated version of itself and then identifying where the absolute value of the output of the correlator reaches its maximum. FIG. 3 is a block diagram of a conventional correlator that can be used for this purpose. A received signal, r(n) is supplied directly to one input of a multiplier 301, and also to an input of a delay unit 303. The delay unit 303 delays the signal by an amount, Tu (where Tu is the duration of the information carrying part of one symbol). In the discussion which follows, N is a number of samples associated with the duration Tu. Typically, N may be the number of samples corresponding to the duration Tu, where N is equivalent to the size of the FFT. It should be noted, however that the invention is not limited to that particular case. The complex conjugate of the output of the delay unit 303 is formed (denoted by the “*” in FIG. 3), and supplied to another input of the multiplier 301. The product (denoted y(n)) generated at the output of the multiplier 301 is supplied to a summing unit 305, which generates a moving sum total of the products. The moving sum represents the amount of correlation, denoted “corr(n)”, which mathematically can be represented by
                              corr          ⁡                      (            n            )                          =                              ∑                          k              =              0                                      NUM_TERMS              -              1                                ⁢                      y            ⁡                          (                              n                -                k                            )                                                                        =                                    ∑                              k                =                0                                            NUM_TERMS                -                1                                      ⁢                          r              ⁢                                                (                                      n                    -                    k                                    )                                ·                                  r                  *                                            ⁢                              (                                  n                  -                  k                  -                  N                                )                                                    ,            where r*(n−k−N) is the complex conjugate of r(n−k−N), and NUM_TERMS is the number of terms in the moving sum.
The phase of the complex valued correlation term, corr(n), can be used to determine the frequency offset. To determine the point at which maximum correlation is reached, the output of the summing unit 305 is supplied to an absolute value unit 307, whose output indicates the magnitude of the correlation value, |corr(n)|.
The result of the complex conjugation and multiplication, y(n), will appear as random noise except when r(n−N) contains the GI and r(n) contains the data copied into the GI. FIG. 4 is a timing diagram that illustrates the relationship between the received signal, r(n), a delayed signal r(n−N), and the moving sum, |corr(n)| for an ideal situation in which the channel has no associated delay spreading.
As can be seen in FIG. 4, if the information carrying part of the signal starts at t=0, the correlation peak occurs at t=−TG. Consequently, for the case in which the peak occurs exactly where expected and Tm=0, one could decide to place the start of the Fast Fourier Transform (FFT) window at the point where the peak is found, or one might alternatively decide to take the start of the window as much as TG later. In practice, depending on how the error in the peak location manifests itself, one should add a certain bias, TB, to the position where the correlation peak is found in order to avoid positioning the FFT window too early. A natural choice for TB is TG/2, since this gives the largest margin for error (i.e., to avoid starting the FFT window outside of the GI).
In case the channel is time-dispersive, the output of the correlator will not show a distinct peak, but rather show up as a plateau. This is illustrated in FIG. 5, which is a timing diagram that illustrates the relationship between the received signal, r(n), a delayed signal r(n−N), and the moving sum, |corr(n)| for a situation in which the channel has a moderate amount of delay spreading.
Again, suppose the information part of the OFDM starts at t=0. If the channel has a maximum delay spread, Tm, the requirement on the start of the FFT window is given by−TG+Tm≦t≦0.  (1)
Thus, as long as Tm≦TG it is possible to avoid ISI if t is chosen according to equation (1). However, if Tm>TG the issue is to choose t such that the effect of ISI is minimized. For systems designed for use in a SFN, the guard interval is typically so large that the first situation is the likelier one.
The time dispersive channel has the effect of delaying the location of the correlator peak compared to the non-dispersive situation. Moreover, the variance of the peak position will increase significantly. The situation becomes even worse in SFNs, where the impulse response of the channel might consist of rays coming from two transmitters which are synchronized, but at very different distances from the receiver. Suppose that the delay spread for the channels between one transmitter and the receiver is small in comparison to the total delay spread experienced by the receiver. The channel might then be modeled as a two ray channel, where the distance between the rays causes a delay spread equal to Tm. It was observed in A. Palin and J. Rinne, “Enhanced symbol synchronization method for OFDM system in SFN channels,” Globecom'98, Sydney, pp. 3238-3243, 1998 (henceforth “Palin and Rinne”), that for such a channel synchronization based on the peak position of the correlator output will not work well. Specifically, if the timing is based on the peak of the correlator, the maximum delay spread that can be handled by the system will be reduced to Tm=TG/2.
The problem was addressed in Palin and Rinne by using two correlators, the second of which has a delay that equals the length of an entire OFDM symbol including the GI. The output from the first correlator is fed to another correlator, and the output from this latter correlator shows a more distinct peak than the output from the first one. If one assumes that the peak will be found in the middle of the above mentioned plateau, that is, at −TG+Tm/2, then it is possible to choose TB=TG−Tm/2. Clearly, assuming that Tm is TG will always give a sampling time that is ISI free. In terms of complexity, however, this approach is much worse since it requires one more correlator, with a delay that equals the length of an entire OFDM symbol including the GI. In addition, in case Tm is significantly smaller than TG, the sampling point will be found close to t=−TG/2 rather than at t=0. Although this will guarantee ISI free reception, it will put unnecessary hard requirements on the channel estimation in the receiver.
Consequently, there is a need to achieve coarse synchronization using an algorithm that is feasible for both small and large values of Tm, and which is not computationally complex.